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Models for RF transformers with extreme ratios?

Discussion in 'Electronic Design' started by Joerg, Apr 29, 2008.

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  1. Joerg

    Joerg Guest

    Hello Folks,

    Is there any model, research results etc. for RF transformers that
    feature extreme turns ratios such 100:1 and more? I am mainly interested
    in leakage inductance, bandwidth and such. Bandwidth doesn't have to be
    more than an octave, single digit MHz range. It just can't be resonant,
    at least not a lot.

    I know this is a far stretch but maybe ...
     
  2. Joerg

    Joerg Guest

    Sorry, typo in one NG address.
     
  3. Tom Bruhns

    Tom Bruhns Guest

    On Apr 29, 11:31 am, Joerg <>
    wrote:
    ....
    ....
    What impedances?

    Try searching for current-sense applications (where the primary is a
    single turn).

    A reasonable starting point for a broadband transformer is a primary
    inductance whose reactance is the same as the primary side impedance.
    So if the bottom end is 2MHz and the primary is designed for 1 ohm,
    you'd want about 80nH or a bit more. An FT114-61 should give you
    about that with a single turn. But 100 turns gives you 800uH which
    will resonate with only 2pF at 4MHz. Properly loaded (10k ohms
    resistive) it should be reasonably damped. A Spice model will tell
    you pretty nicely what the response will be for any reasonable assumed
    coefficient of coupling; for example, assuming 80nH:800uH with k=0.9
    (which seems like it ought to be pretty easy), and 2pF||10k secondary
    load, driven from a 1 ohm source, you get a 3dB bandwidth from about
    1MHz to 12MHz, with no apparent resonance effects. k=0.8 and
    Cload=4pF only cuts the top end to about 6MHz. (The bottom is
    determined mainly by the reactance. Get to low enough coupling and
    the effective turns ratio drops, but at k=0.8, the mid-band is only a
    fraction of a dB below the "theoretical" 40dB voltage stepup.)
     
  4. Joerg

    Joerg Guest

    About what you assumed, around an ohm at roughly rectified AC level.
    That one ohm will be a whole 'nother story but I'll get it there.
    Somehow :)


    Thanks, Tom. I am currently at around 300-400nH primary but any
    capacitance on the stepped up side is killing things. The secondary in
    those cases is always largish because it needs to withstand a lot of
    breakdown voltage. The only feasible method is to wind the packet on top
    of the primary or use compartmentalized bobbins and make sure the other
    layer maintains enough clearance. The latter not so much for HV
    breakdown but to avoid stray capacitance to ground.

    Probably the best avenue is to get a suitable large core and make one. I
    was hoping there was some example data from core/bobbin manufacturers
    and such but the usual suspects didn't have anything.
     
  5. Tom Bruhns

    Tom Bruhns Guest

    Oh, yeah, and I forgot to ask: how much power? ;-) My comments came
    from a low-power context, though they tranlate. The
    reactance:impedance thing should stay the same, assuming you avoid
    core nonlinearity. It actually came as a bit of a surprise to me how
    low a winding reactance is when you get to the low frequency cutoff.
    The relatively simple model I suggested has worked well for me: get
    the coupling coefficient up to extend the high end. The model matches
    several RF transformers I've measured. Up till resonances and other
    capacitive effects get to you, you can generally extend the response
    of a transformer by driving it with a lower source impedance and
    loading it with a higher impedance. I have a 1:1 audio transformer
    that, when driven with a low impedance and loaded with about 2k ohms,
    is flat within +/-0.1dB from 0.6Hz to 109kHz, but quite a bit worse if
    driven from 600/loaded with 600, and it shows resonant peaking at the
    high end if loaded too lightly.

    Suggest you go for a core with modest permeability, probably around
    100 (depending on path area and length), so you can drop the
    inductance down some from where you are. Harry D. seems to know a lot
    about this sort of thing; maybe he'll have some ideas.

    Cheers,
    Tom
     
  6. Eric Tart Red "Arbeitsbush fur den HF Techniker"

    The very first chapter is about the RF transformers and the simulated
    line transformers, their equvalents and the compensation.

    But the ratio of 100:1 in one stage doesn't seem reasonable; the
    sensible way would be breaking this into the series/parallel connection
    of the transformers.

    This type of the impeadance matching can be done by a bandpass filter
    type of network (also using many stages); I would do it that way.


    Vladimir Vassilevsky
    DSP and Mixed Signal Design Consultant
    http://www.abvolt.com
     
  7. Joerg

    Joerg Guest


    Not at liberty to tell ;-)


    Yes, it's a compromise to push the upper end a bit. On line transformers
    it's to save cost on the copper. So when I need really low standby power
    I often use a 230V transformer at 120V.

    Yes, a really low drive impedance is key. I'll just whip up a few and
    measure them on the bench. The data I can find at manufacturers is
    mostly based on usage at the impedance they are marketed for.
     
  8. Joerg

    Joerg Guest

    On extreme ratios you just can't avoid it.

    Yes, and you can never have half turns like at some restaurants.

    Unfortunately that won't work in this case. It's usually only ok if you
    have an active stage inbetween.

    Maybe you could come visit at the hospital if it blows up in my face ;-)
     
  9. Joerg

    Joerg Guest

    As usual it depends :)

    A larger primary inductance reduces core losses, it thus becomes less
    warm and under light loads it can be a good thing. At higher currents
    copper losses would cause a penalty.

    Then main advantage is that the core won't saturate if, say, a unit is
    running on generator power and an impatient maintenance guy gooses the
    throttle a bit until the old spark plugs that should have been replaced
    last year burn themselves cleaner and it stops misfiring.
     
  10. Joerg

    Joerg Guest

    Not quite that bad but yes, you need to provide a bigger one. For high
    power you'd be better off buying or designing a good 90VAC-260VAC switcher.

    I've seen a lot of grief in that area :-(

    A lot of engineers believe all that can be permanently muffled by some
    MOV here and there. However, those work like a bank account and one fine
    day ... KABLOUIE.
     
  11. Tom Bruhns

    Tom Bruhns Guest

    Oh, ye of little faith. I have some FT82-67 toroid cores. I wound
    one with 100 turns of #29AWG solid enameled copper wire, and put one
    turn of #14AWG through as a primary. I loaded the secondary with 10k
    ohms in series with 51.1 ohms, with the receiver port of an HP8753E
    across that 51.1 ohms. The primary is in parallel with 1 ohm, and 50
    ohms goes from that off to the source port of the analyzer. The
    response below 20MHz is what I consider to be very close to what my
    model predicts; I get a peak response at 10.68MHz, with bandwidths
    -0.5dB 7.28-12.71MHz, -1.0dB 5.72-14.24MHz, -2.0dB 3.48-16.69MHz and
    -3dB 2.89-18.76MHz. I would expect that Joerg, at his much higher
    power levels, should still have relatively little trouble getting an
    octave bandwidth in a single transformer (to -1dB, anyway) at the
    somewhat lower frequencies he's dealing with. He admittedly will have
    to be careful to avoid actual power loss and to minimize stray
    capacitance.

    Cheers,
    Tom
     
  12. amdx

    amdx Guest

    Please give some more information about why you can never have half turns.
    I remember
    overheating a transformer that used a half turn. Never tried half turns
    again. But what is the reason the half turn gets hot?
    Mike
     
  13. Tom Bruhns

    Tom Bruhns Guest

    With a toroid core, it should be obvious why you can have only
    integral numbers of turns. In an E-I core, or a pot core with
    openings on both sides, you can have a wire exit a different place
    than it entered. The loop then closes around one of the outside
    "legs" of the core. Note that this is equivalent to a full number of
    turns around the center post, and one turn around the outer post, with
    the two connected in series. IF the magnetics are balanced, the field
    in the outer leg will be half the field in the center leg. But this
    happens only if there is no current in the turn around the outer leg.
    Note that the "half turn" is not strongly coupled to the rest of the
    turns, and as a result adds a lot of leakage inductance. I don't see
    why the "half" turn itself should get hot, but if it diverts the
    magnetic field into the other leg in such a way that it significantly
    increases the core loss in that leg, it could lead to excess power
    loss in the transformer.

    Cheers,
    Tom
     
  14. Tom Bruhns

    Tom Bruhns Guest

    Which got me to thinking: you can keep the magnetics in the two outer
    legs balanced (that is, the rate of change of flux per unit time) if
    you put a turn around each and put those two turns in parallel.
    However, each will see half the flux that's in the center leg, so will
    contribute half a turn's voltage...this could be an interesting way to
    get a high step-up ratio with fewer secondary turns: only 50 turns
    instead of 100, to get a 1:100. That could be an advantage in keeping
    the parasitic capacitance on the secondary at bay, though the
    effective capacitance is generally a very weak function of the actual
    number of turns--and for modest permeability cores, significantly
    lowers the pri:sec coupling as compared with having the windings co-
    axial.
     
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