Connect with us

CPLD Jitter

Discussion in 'Electronic Design' started by Andrew Holme, Aug 28, 2005.

Scroll to continue with content
  1. Andrew Holme

    Andrew Holme Guest

    The dividers and the phase detector of my experimental frequency synthesizer
    are implemented in a 15ns Altera MAX7000S CPLD. I've tried different
    multiplication factors (kN) to see how the close-in phase noise varies. At
    a 1 KHz offset, I get:

    -82 dBc/Hz for N=198 (VCO=19.8 MHz, comparison freq = 100 KHz)
    -95 dBc/Hz for N=39 (VCO=19.5 MHz, comparison freq = 500 KHz)

    Calculating the equivalent phase noise at the PFD:

    -82-20*log10(198) = -128 dBc/Hz
    -95-20*log10(39) = -127 dBc/Hz

    Given the 5:1 ratio of comparison frequencies, at a guess, I'd expect these
    to differ by 13 dB if the noise was mainly due to a fixed amount of time
    jitter at the PFD.

    I'm using a 10 MHz canned crystal oscillator, which I'm dividing down
    (inside the CPLD) to obtain the reference frequencies. I've read these are
    good for at least -130 dBc/Hz (before dividing down) so I'm a bit
    dissappointed with my noise levels. Maybe it got a bit too hot when I
    soldered it to the ground plane! I must try another....

    Googling for "altera cpld jitter" doesn't turn-up much, and they don't
    mention jitter in the datasheet. Does anyone know what sort of performance
    can be expected from a CPLD in this regard? I don't know if the CPLD, or my
    circuit lash-up is the root cause.

    A full write-up of the project can be found at
    http://www.holmea.demon.co.uk/Frac2/Main.htm It has a fractional-N
    capability, but noise-levels are the same in integer-N mode with the
    external RAM disabled.

    Thanks,
    Andrew.
     
  2. Ken Smith

    Ken Smith Guest

    If you have other signals running through the CPLD, part of the problem
    could be crosstalk. Power supply noise will also show up as a jitter.

    There seems to be a bad solder joint in the upper left corner.

    Check the LM78M05 for oscillations. We don't see the traces hooking up the
    bypasses on it.


    You also have to be careful when making things like flip-flop phase
    comparitors inside the MAX series. When I made one, I had to add some
    extra logic after the flip-flops so that the "both" state of the flip-flop
    pair was used to gate off the output. I think the CLRN timing was the
    problem bit once I got it working, I stopped looking for the root cause.

    IIRC:

    BothLatch = (UpFlipFlop # DownFlipFlop) & BothLatch
    # UpFlipFlop & DownFlipFlop;

    Up = UpFlipFlop & !DownFlipFlop & !BothLatch;
     
  3. Hal Murray

    Hal Murray Guest

    Can you measure the raw oscillator output?

    It may not be as good as you expect. Many low volume oscillators
    now use a PLL. They program the dividers rather than grind the crystal
    to get custom frequencies.

    I'd expect that numbers like 10 MHz would have enough volume so that
    they would avoid the PLL but maybe it's cheaper to have one production
    setup and use it for the common frequencies too.
     
  4. Daniel Lang

    Daniel Lang Guest

    I would suggest inserting a 100 ohm resistor between C3 and C7 and placing
    a 100uF capacitor in parallel with C7. The noise on the output of U4 is
    just as critical as the noise on the + input of U3.

    You may also want to add an RC filter on the output of U5 before feeding
    it to U6.

    Daniel Lang
     
  5. Joerg

    Joerg Guest

    Hello Andrew,
    Check its data sheet. If it doesn't have any noise specs on there get a
    better one. Or better yet, roll your own. CTS is a good source though.
    How are the Altera GND pins connected to the plane? I can't see any
    connections. If that is via traces that's the first easy thing to fix.

    Look at your VCC. Switch the scope to AC and crank it way up. When
    called out to clients that is were I found the main contributors for
    phase noise. Video line sync, RAM banking spikes, whatever, it all ended
    up adding a little AM modulation to the logic output stages which in
    turn causes jitter.

    Sometimes others thought now I'd gone crazy: It often helps to take a
    very good comm receiver, put on tight fitting headphones and listen to
    your signals. Both sidebands plus wide open w/o IF filter. Often that
    revealed the tiny telltale "weeeee" or "rat-tat-tat". A spectrum
    analyzer on the pins and also on VCC can help as well.

    Then there is the usual, such as 0.01uF caps tightly fitted from VCC
    pins (all of them) to the ground plane. Also, check what else is
    happening in the FPGA. As Ken wrote this could cause crosstalk,
    especially if other outputs are heavily loaded or must drive longer traces.

    Regards, Joerg
     
  6. Brian Davis

    Brian Davis Guest

    Andrew,
    Asides from the supply/clocking suggestions already made,
    I'd also add the following caution: the output stages of most
    programmable digital parts are not intended for producing
    clean reference signals to a PLL used for low-noise RF signal
    generation.

    A few times over the years that I've considered doing this,
    the first thing I've done is build something like a pulse output
    divide-by-ten in the intended technology, driven the part with an
    oscillator having known phase noise, and looked at the divider output
    on both a spectrum analyzer & phase noise measurement system.

    Most of the programmable parts I've measured have had output spurs
    50-60 db down that vary strongly in frequency with supply voltage,
    and can cross over the intended output frequency, making it impossible
    to keep them out of the loop BW. ( also, the crud in the output
    spectrum will generally behave differently with an even duty cycle
    output than for a pulse output divider )

    other (hastily conceived) random thoughts:
    - what's the noise floor of your spectrum analyzer at 1 KHz?
    - try using a surplus ocxo for your 10 MHz source
    - if you disable the uC/SRAM, and implement a fixed divide entirely
    internal to the CPLD, does the noise get any better?

    have fun,
    Brian
     
  7. Andrew Holme

    Andrew Holme Guest

    Thanks to all for the suggestions.

    The ground pins were connected by threading wires through the holes, and
    soldering them to the ground plane before fitting the PLCC socket.

    Personally, I now think the monolithic regulator supplying U4 is the main
    noise source. Thanks, Daniel. I've been pouring over Rohde and Egan,
    trying to get my head round spectral density stuff. I'd appreciate it if
    someone could check my math. I've calculated the equivalent noise voltage
    that would have to be injected at the loop filter input (U4 output) to
    produce -95 dBc/Hz at a 1 KHz offset on the VCO. The following script was
    executed in SCILAB:

    // Closed-loop gain from PFD output to VCO output
    pd_2_vco = h/kpd/kn;

    // -95 dBc/Hz
    theta_rms = sqrt(10^(-95/10) * 2);

    // Equivalent noise injection at PFD output (1 KHz offset)
    v_rms = theta_rms / horner(pd_2_vco, 2*%pi*1000)

    This gives 20nV/sqrt(Hz). Since U4 output is a 50% duty cycle square
    wave(XOR PFD), presumably I would still only need 40nV/sqrt(Hz) on the
    regulator output? Is 40nV/sqrt(Hz) at 1 KHz credible for a 78L05 with 100n
    + 10n hanging off its output?

    Script notes:
    h = Closed loop gain
    kpd = PFD gain
    kn = Divider gain
    horner() returns magnitude of transfer function at specified freq
     
  8. Andrew Holme

    Andrew Holme Guest

    The 78L05 datasheet quotes an output noise voltage of 40uV for 10Hz <= f <=
    100 KHz with a minimum recommended load capacitance of 10n. I presume this
    means 40uV peak-to-peak? I'm not sure how to convert this to RMS
    nV/sqrt(Hz), but 40e-6/sqrt(100e3) = 1.26e-7 which is only 3 times my
    figure.
     
  9. Andrew Holme

    Andrew Holme Guest

    Sorry, I think that was wrong. That h was the closed-loop gain from ref
    input to vco output.

    A safer way to calculate it, using the SCILAB /. operator is:

    t_pd_vco = (f * kvco/s) /. (kpd * kn);

    where f = loop filter transfer function.

    Now I get v_rms = 793 nV/sqrt(Hz)

    Hmm....
     
  10. Andrew Holme wrote...
    Hah, it's likely rms, because that leads to a much smaller number.
    Also, the NEC and Linfinity datasheets explicitly say rms.

    The NSC datasheet note says, "minimum load capacitance of 0.01µF
    to limit high frequency noise," which means the output noise is
    not white, which means you can't perform the usual simple sqrt-BW
    calculations to obtain the noise density.
     
  11. Andrew Holme

    Andrew Holme Guest

    Mike, thank you. I just had one of those "Doh! Why didn't I think of
    that?" moments. I think you may have solved the mystery. The loop
    bandwidth is only slightly wider than 1 KHz, so of course there is very
    little attenuation of VCO noise at that frequency! I've just checked
    the predicted response of my loop for noise injected by the VCO, using
    my SCILAB model, and it agrees to within a few dB with the
    measurements. I'd like it to be closer though .....
     
Ask a Question
Want to reply to this thread or ask your own question?
You'll need to choose a username for the site, which only take a couple of moments (here). After that, you can post your question and our members will help you out.
Electronics Point Logo
Continue to site
Quote of the day

-